Magnetic resonance gyro signal processor

ABSTRACT

A signal processor which involves separation of the cell signals from a magnetic resonance gyro (MRG) using phase locked loops that also serve as frequency multipliers, phase shifting one cell output to maintain phase comparisons within the monotonic region of the phase detectors, PDM/Digital conversion of the MRG signals and finally microcomputer processing to obtain gyro angle to the required resolution and update rate.

CROSS-REFERENCE TO RELATED APPLICATIONS

This is a continuation-in-part of application Ser. No. 931,702, filedAug. 7, 1978 abandoned.

PRIOR ART

U.S. Pat. No. 3,778,700: Bayley et al, December, 1973

U.S. Pat. No. 3,808,542: Ferriss, April, 1974

"Picture Coding Using Pseudo-Random Noise" by Lawrence Gilman Roberts,appearing in the February, 1962 IRE Transactions on Information Theory(IT-8, No. 2, pp. 145-154, February, 1962).

"Delta Modulation Systems" by R. Steele, pp. 296-299, published in 1975by Halsted Press, a Division of John Wiley & Sons, Inc., N.Y.

"Dither Signals and Their Effect on Quantization Noise" by LeonardSchuchman, appearing in IEEE Trans. Commun. Technol., vol. COM-12, pp.162-165, December 1964.

"The Application of Dither to the Quantization of Speech Signals" by N.S. Jayant and L. R. Rabiner, published in American Telephone andTelegraph Company, The Bell System Technical Journal, Vol. 51, No. 6,July-August, 1972.

This invention is related to the magnetic resonance gyro. Moreparticularly, this invention is a signal processor for converting thegyro output angle to digital form.

BACKGROUND OF THE INVENTION

The atomic nuclei of certain materials possess magnetic moments whicharise out of their inherent angular moments or spin properties. Thesespecial properties of certain materials form the principle upon whichthe nuclear gyro operates.

U.S. Pat. No. 3,778,700, assigned to the same assignee as the presentinvention, discloses an optically pumped nuclear magnetic resonancegyroscope device in the preferred embodiment of which two differentisotopes of mercury are utilized in the cells of two interconnected spingenerators. As taught therein, each sping generator comprises a mercuryabsorption cell containing ¹⁹⁹ Hg and ²⁰¹ Hg which is subjected to a DCmagnetic field, H_(o) and to an AC magnetic field, H₁ perpendicular tofield H_(o). The magnetic fields H_(o) of the two cells are described asbeing equal and antiparallel to each other. The mercury absorption cellsare optically pumped by a circularly polarized beam of light to increasethe magnitude of the net magnetic moment and phase comparison means areprovided to derive a readout signal representative of gyro angularrotation about a given axis by comparing the phase outputs of the twospin generators.

The disadvantages of the prior art early schemes were:

1. Gyro output was in analog forms not readily useable by modern-daynavigation computers without conversion.

2. Mechanization required bulky and expensive electro-mechanical deviceswhich are slow in response time.

3. Stability of the phase shift introduced by the resolver scheme isdependent not only on the resolver characteristics, but also on thestability of the resistor and capacitor (the phase shifter works ideallyonly at a single frequency).

4. Filters prior to the resolver introduce phase instability with timedegrading output.

Thus, there is a need to implement a system which will provide digitalsignals which will satisfy the fast midcourse corrections of present daymissiles and other gyro controlled devices.

BRIEF SUMMARY OF THE INVENTION

Briefly, the apparatus of the present invention comprises a magneticresonance gyro having a digital phase shifter which converts the gyro'sinherent analog signals into digital pulse count and a residual PDMsignal which is further converted to a digital word. The set ofdigitized signals are applied to a microprocessor system where thedigital gyro output angle is constructed for use by the navigation orguidance computer.

Accordingly, it is an object of this invention to provide a means forconverting the analog signals from a magnetic resonance gyro to digitalsignals for direct measurement of gyro angle by modern-day navigationand guidance computers.

This and other objects, features and advantages of the present inventionwill become more apparent from the following description taken inconjunction with the accompanying drawings wherein:

FIG. 1 is a simplified block diagram of a conventional magneticresonance gyro;

FIG. 2 is a block diagram in servo-oriented format of the mechanizationof the gyro elaborated upon by this invention;

FIG. 3 is a more detailed block diagram of the phase shifter of FIG. 2;

FIG. 4 shows the signal inputs to the phase shifter of FIG. 3;

FIG. 5 shows the selected monotonic range of the transfercharacteristics of the phase detector of FIG. 2;

FIG. 5A shows a block diagram of the PDM modulator;

FIG. 5B is a graph showing how the actual gyro angle is recovered whenit falls between PDM modulation levels;

FIG. 6 shows the pulse duration modulation to digital converter;

FIG. 7 shows the three full periods of the pulse duration modulationinput waveforms and the corresponding digital output; and

FIG. 8 shows the microprocessor with the outputs of the PDM/Digitalconverter and with the Δφ pulses inputted to an Up/Down counter.

Referring to FIG. 1, there is shown a simplified block diagram of amagnetic resonance gyro that outputs an analog signal which issubsequently digitized according to the invention. The spin generatorcells 10 and 11 comprise absorption cells containing ¹⁹⁹ Hg and ²⁰¹ Hgwhich is subjected to a DC field H_(o). In the spin generator loops, thephase stable amplifiers 12 and 13, receive input signals fromphotodiodes (not shown) associated with each cell and also supply energyto the nuclei of the mercury isotopes to sustain steady-state Larmoroscillations. The composite cell signals are supplied to the signalprocessor 14 which provides angular output information.

A more detailed diagram in the classical control system format of amagnetic resonance gyro mechanization is shown in FIG. 2. Cells 10 and11 are shown in dotted outline. The four phase locked loop (PLL)frequency/phase multipliers 29, 30, 40 and 42 scale output phases by afactor N, prior to phase comparison in phase shifters 31 and 32. Becausephase detectors 37 and 39 are monotonic over a range of only π radianssome kind of phase shifter is required to prevent sense changes. Theneed for a phase shifter can be recognized by assuming a fixed inputrate considering the angles which would accummulate at the inputs to thephase detector over time. The phase shifter is introduced by GΔφ block36 which feeds back a phase identified as MΔφ tending to restrict themagnitude of the output of the phase detector. In earliermechanizations, this was implemented using a pair of resolvers for thephase shifters with shafts connected together. The GΔφ block 36 was aservo amplifier driving a motor to which the resolvers' shafts wereconnected. Thus the output Δθ₂ of phase detector 39 is driven to null bythe null-seeking servo containing the GΔφ block 36. Under thatmechanization, the integrated rate, ωn/ S=θ_(r), is given by:

    θ.sub.r =-MΔφ/2N                           (1)

Thus, the gyro output is proportional to the resolver shaft angle andmay be outputted in this manner. Requirements for strapped-down gyrosfor use in navigation systems in the 1 n.m. per hour class call for anangular resolution of 1 sec. To operate a resolver/synchro output todigitize to that resolution it would require a 20 bit converter toachieve 1.24 sec. resolution: ##EQU1## This resolution requirementsurpasses the performance capabilities of fast rate tracking converters.

The essentials of the digital phase shifter of the present invention canbe explained by using FIG. 2. Here the output of phase detector 39 isallowed to accumulate an output angle Δθ₂ by placing a dead-band inblock 36 (GΔφ). When Δθ₂ exceeds the dead-band a quantum of phase shiftΔφ is introduced into phase shifters 31 and 32 within the period of thephase detection, thus shifting Δθ₂ by an amount -Δφ. The number of Δφ'sintroduced is accumulated in a counter (see reference symbol 82 in FIG.8) which totalizes Δφ pulses thus containing ±MΔφ. The total output ofthe gyro is contained by the quantized value MΔφ and the residual Δφ₂.Moreover, block 34 (GΔH) is operated as a null-seeking servo forcing Δφ₁=Δφ₂. The quantities labelled OUTPUT are:

    Δφ.sub.1 =2N.sub.θr +MΔφ         (3a)

    Δφ.sub.2 =-2N.sub.θr -MΔφand Δφ(3b)

In the PDM to digital converter (FIG. 6) digital conversions of Δθ₁ andΔθ₂ are performed at a binary submultiple level of Δφ. The remainingoperations include subtraction of (3b) from (3a), subtraction of MΔφfrom the result and division by 4N by bit location recognition, thusproducing θ_(r), all performed by the microcomputer of FIG. 8.

Having briefly described the overall signal processing scheme, each ofthe elements will now be considered in more detail.

Turning to FIG. 3, a block diagram of the digital phase shifter 31 isshown. Block diagram 32 is identical to block 31 and the descriptionwhich follows applies equally to both. The two input signals, Nθ_(2A)and Nθ_(2B) to the phase shifters 31 and 32 are square waves from thephase-locked-loop frequency multipliers which have raised the celloutput signals to a frequency N times the original. Input signal Nθ_(2A)is applied along with clock pulse CL2A to eight stage shift register 53and input pulse signal Nθ_(2B) along with clock pulse CL2B is applied toadvancer retarder detector 54. The clock pulse inputs are square wavesat a frequency eight times the input signals. These signals aregenerated within the phase-locked-loop frequency multipliers and arefrequency and phase-locked with their respective input signals as shownin FIG. 4. The output from the eight stage register 53 is applied toeight channel multiplexer 56. Also applied to block 56 are three bitaddress lines from three stage up/down counter 57. The advancer retarderdetector provides an output signal +Δφ and -Δφ which are applied togating circuit 58 and to the angle counter (not shown). Gating circuit58 has a third input, clock pulse CL2A and outputs to up/down counter57. The output of the eight channel multiplexer 56 is a signal Nθ_(2A)-MΔφ which is applied as an output signal to the phase detector 59 andto the advancer retarder detector 54.

The fundamental purpose of the digital phase shifter is to maintain thesignals Nθ_(2A) -MΔφ and Nθ_(2B) which go to the phase detector within aselected monotonic range of the transfer characteristic as shown in FIG.5 for an Exclusive OR gate 59. The output can be in terms of either thelogic "1" level output duty cycle or the average value of the outputvoltage. The point to be made is that the transfer characteristic isperiodic having positive and negative slopes. The region to whichoperation is confined by the action of this digital phase shifter isbetween π/8 and 7π/8, shown by the dotted lines. Within theadvancer-retarder detector 54 an electronic "window" is effectivelygenerated. Relative phases Nθ_(2A) -MΔφ and Nθ_(2B) which fall withinthis window produce no activation of the phase shifter. As the relativephase equals or exceeds the window limits an output pulse is generatedon one of the lines +Δφ or -Δφ. After synchronization to the CL2A withingating circuit 58, three stage ring counter 57 is incremented ordecremented. The counter contents represented by the states of threeflip-flops determine a binary number 0 through 7 which representsclosure of one of eight input switches of multiplexer 56. Theincrementing or decrementing of the ring counter 57 causes the switchpreviously closed to open and an adjacent switch to close which connectsthe correspondingly numbered state of shift register 53 to the outputline of multiplexer 56. Within shift register 53 at any time are eightimages of the input Nθ_(2A) each shifted in phase by π/4 (1/8 of 2π) bythe action of the clock CL2A operating at eight times the inputfrequency. The selection of an adjacent stage of shift register 53 thusis tantamount to shifting the output phase by a quantized increment ofπ/4 radians. The appearance of Nθ_(2A) -MΔφ on the output line, with Mnow having been incremented or decremented by one, effects the phaseshift of π/4 causing the phase comparison, previously residing at eitherπ/8 or 7π/8 on the phase transfer characteristic to shift toward thecenter of the characteristic, to either 3π/8 or 5π/8 dependent on phaseadvancement or retardation. The ±Δφ pulse generated is accumulated inthe angle counter of FIG. 8 as a measure of total digital phase shift,±MΔφ.

The completion of the phase shifting sequence is achieved within threeclock pulses allowing a maximum phase shifting rate of one quantum perperiod of input. That is, dm/dt is equal to the frequency of the input:

    dm/dt=Nf.sub.c                                             (4)

where f_(c) is the frequency of Larmor oscillation associated with γ₂.

Differentiating equation (1) with respect to time and substitutingequation (4): ##EQU2## For Δφ=π/4, f_(c) =1000, ω_(r) =12.5π or greaterthan 6 revolutions per second. This is the only electronics-imposed ratelimitation apparent at this time.

Turning to FIG. 6, there is shown a schematic block diagram of the PDMto digital converter. A PDM input from the phase modulator of FIG. 5A isapplied to phase detector 60 and gating circuit 61. Phase detector 60also receives a signal from counter divider 62. The output of phasedetector 60 is applied to low pass filter circuit 63 and the output ofcircuit 63 is applied to voltage control oscillator 66 which convertsthe dc input into pulse output. The blocks enclosed in the dottedoutline are a conventional PLL (phase-locked loop) frequency multiplier.The pulse signals from the voltage control oscillator 66 is applied tothe counter divider 62 and gating circuit 61. Gating circuit 61 providesfour output signals for application to counter 67. Gating circuit 61provides serial output signals on line c and counter 67 providesparallel binary word output. The output frequency is P times the inputfrequency f_(in). Or another way of thinking of it, within each periodof the input are generated P output pulses. Usually a binary ripplecounter is used as the frequency divider such that P is some integralpower of 2. U.S. Pat. No. 3,808,542 teaches a PDM to digital converterwhich may be employed in the invention.

The total gyro angle θ_(r) is contained in the pulse count M multipliedby the phase shift weighting of Δφ plus the residual angle representedby the index of modulation of the pulse duration modulation outputs ofthe phase detectors.

When 50% duty-cycle square waves are applied to the exclusive OR gate 59(FIG. 5) used as a phase detector, the output signal is a PDM wave at afrequency twice that of the input frequency. The range from -100%modulation (duty cycle=0) to +100% modulation (duty cycle=maximum)corresponds to a range of the phase detector transfer characteristic ofπ (r). From equation (3) that equivalent gyro angle is:

    π=2Nθ.sub.p

The equivalent gyro angle stored in the PDM wave is:

    θ.sub.p =π/2N (rad)                               (6)

By operating the multiplied input frequency and the PDM input signalinto AND gate 61 an output pulse train is generated wherein for eachperiod of the input PDM wave the number of output pulses, expressed as afraction of P is numerically equal to the duty cycle of the input PDMwaveform. By use of gating circuit 61 the counter 67, with presetcapability and built-in latches, can output in the following manner:

1. During that portion of the PDM input where the input is high or atthe logic 1 value the binary output word is that from the previous PDMperiod.

2. At the point in time where the PDM input goes low the binary outputchanges to a value representative of the current PDM period.

This is illustrated in FIG. 7 where three full periods of a PDM inputwave are shown with the duty cycle varying linearly from 5/8 to 3/8.Assume for example P=2⁸ =256. Since the value of the PDM wave precedingthe first shown, the output value is unknown except that it would liebetween 00000000 and 11111111. At the negative going transition of thefirst PDM wave K=0.625×256=160 pulses would have incremented the counterto read 10100000 and the output would be latched at that time, afterunlatching the counter is zeroed to await the positive going input ofthe next PDM wave. Meanwhile, the latches cause the output to remaindisplayed until updating at the next negative going transition.

Now, fixing more numbers to this example, assume N=8 for the multipliersassociated with γ₂ frequency of 1 KHZ. By equation (6) the PDM wave, at16 KHZ, contains 11.25° of the equivalent input angle. With P=2⁸ theleast significant bit of the contents of the counter in the PDM toDigital Converter (or the angular value of the serial pulse) is11.25°÷2⁸ =0.044. The frequency of operation of the VCO in the converteris 16 KHZ×2⁸ =4.096 MHz.

Other options are available using this general scheme. The counter couldbe present to -P/2 instead of being zeroed thus removing the offsetangle. Also, multiple period summations could be generated by zeroing orpresetting and latching after a selected number of PDM periods havetaken place. If the number A is chosen as the number of periods overwhich summation takes place, equations for the quantization level andthe time between updates can be developed and are: ##EQU3##

Turning to FIG. 5A PDM modulator 100 receives PDM signals from phasedetector 59 shown in FIG. 5. Where the technique for improving angularresolution by multiple period summations is used a means of linearlyphase modulating the output of phase modulator 100 is required. Thephase modulating signal may be obtained from the microprocessor of FIG.8. Modulating the PDM signal with a linear modulating signal having apeak-to-peak amplitude at least as large as the PDM quantization levelassures that gyro angles residing between PDM quantization levels aremade apparent to the PDM to digital converter and subsequently to thesuccessive summation. Those skilled in the art of integer quantizationaveraging techniques will recognize that the improved estimate of gyroangle, increased in resolution by a factor A, is biased to a lower valueby an amount equal to one half the quantization level of the PDM todigital converter. However, the bias angle is easily added during thesummation process.

Equation (6) is the gyro angle stored in the PDM wave as modulation:

    θ.sub.p =π/2N (rad)                               (6)

The PDM to digital converter demodulates the PDM wave resolving the gyroangle further by the factor P, the multiplication factor of the phaselocked loop (PLL) in the PDM to digital converter. Thus the quantizationlevel associated with the output of the PDM to digital converter is:##EQU4##

FIG. 5B is a pictorial representation of a case where the actual gyroangle θ_(g) lies between two adjacent quantization levels, shown ashorizontal lines 101 and 102, separated by θ_(m). The gyro angle shownhas a value

    θ.sub.g =θ.sub.o+θ.sub.m +kθ.sub.m (6b)

In the absence of any systematic linear PDM modulation the PDM converterwould output the angle representative of the quantization level justsmaller than the true gyro angle. The maximum error approaches θ_(m) ask approaches 1.

Additional resolution can be obtained by modulating the PDM waveformwith a linear periodic wave and successively summing PDM conversionsamples over a period of the modulation. One period of triangular PDMmodulation (period T₈₂) is shown. The modulation has a peak value θ_(m)and an absolute slope 4θ_(m) /T.sub.μ.

The presence of the modulation causes PDM conversion to quantize atthree levels. The average value of the converted angle is ##EQU5##Comparison with equation (6b) shows this value is less than the actualgyro angle by one half the quantization level. Those skilled in integerquantization averaging recognize that this bias angle of θ_(m) /2 isnormal and expected for this type of signal processing. If A successivesums are accumulated over a period Tμ resolution is again improved bythe factor A.

Referring again to FIG. 5A, the quantization times at low, intermediateand high levels are calculated as follows: ##EQU6##

Since the actual gyro angle is θ_(g) =θ₀ +kθ_(m) +θ_(m) /2, the averagecomputed angle is quantized 1/2 a quantization level less (θ_(m) /2).

The frequency of oscillation of the VCO in the converter is the limitingfrequency and can be expressed by:

    f.sub.(VCO) =2NPF.sub.c(γ)                           (9)

Assuming a center frequency of 16.384 MHZ as a conservation choiceequation (9) produces a value NP=8.192×10³. Imposing a 1.24 secresolution requirement from equation (2) and solving for A in equation(7) a value of 32 is obtained. Finally from equation (8) a time betweenupdates of 2 ms. is obtained. The combination of 1.24 sec resolution, 2ms. update rate and the rate handling capability of 2250 degrees persecond satisfies not only the most stringent requirements for strappeddown navigation systems, but also for all but the high spin ratestabilized tactical guidance applications.

Referring to FIG. 8, the processing of the digital signals isaccomplished by use of the microprocessor shown. The outputs of the twoPDM/Digital converters are inputted to a pair of asynchronous interfaceadapters 80, 81 while the Δφ pulses are counted by an up/down counter 82and paralleled into a peripheral interface adapter 83. Interconnectingthe microprocessor unit with I/O interface and interface I.C.'s(integrated circuits) and memory I.C.'s (integrated circuits) is theparallel data bus 84. The read only memory 86 (ROM) contains the programwhich operates on equation (3) to yield gyro input angle θ_(r) with theappropriate summation. The random access memory 87 (RAM) stores interimcomputations, performed by the microprocessor 88, or in an expandedprocessing version of this microcomputer, stores moving averagecomputations, based on a triangular averaging technique. At the outputof the microprocessor through a pair of peripheral adapters 89 and 90 isthe 20 bit binary word (1.24 sec LSB) representing total MRG gyro inputangle θ_(r), updated at the required rate.

While particular embodiments of the invention have been shown anddescribed, modifications may be made, and it is intended in thefollowing claims to cover the embodiments which fall within the truespirit and scope of the invention.

What is claimed is:
 1. A signal processor operating on inherently analogsignals for converting them to digital words representing gyro anglecomprising:a source of composite analog signals containing gyro angulardisplacement, means for separating and frequency-multiplying said gyroanalog signals retaining phase coherence, phase detector means fordetecting and comparing said frequency multiplied signals and forproducing a residual PDM signal, means for incrementally phase shiftingsaid frequency multiplied signals and accumulating the number of saidincrements such that subsequent phase comparisons remain within a fixedmonotonic region of said phase detection means, means for operating onsaid residual PDM signal producing a digital word representative of theduty cycle, and means for scaling and adding said accumulated number ofsaid increments produced by said phase shifting means to the result ofsaid digital word thereby producing digital output words representinggyro angle, wherein said means for incrementally phase shifting saidsignals comprises: a shift register containing eight images of said gyroanalog signals, a multiplexer connected to said shift register formaintaining the phase shifted signal within a selected monotonic range,an advancer-retarder detector connected to said multiplexer fordeveloping an electronic window for retarding the analog signals whichfall within said electronic window, a gating circuit to maintain phasedetection inputs within a monotonic range, and an up/down counterconnected to said gating circuit and providing a three bit address tothe input of said multiplexer.